High-frequency switching circuits operable in a natural zero-voltage switching mode

ABSTRACT

A dc-to-dc power converter, comprising a transformer, a single primary-side power switching device and a parallel combination of a diode and a capacitance, is controlled to operate in a &#34;natural&#34; zero-voltage switching mode such that the power switching device is switched with zero-voltage thereacross. Due to the simplicity of the circuit, i.e., very few components, and its high-frequency capability, it can be implemented using high-density packaging techniques. Such a converter is useful in ultra-high-density point-of-load dc-to-dc converters for distributed power systems.

This application is a continuation of application Ser. No.08/262,177 filed Jun. 20, 1994 now abandoned.

FIELD OF THE INVENTION

The present invention relates generally to power converters and, more particularly, to dc-to-dc power converters amenable to high-density packaging techniques and very high frequency operation, which converters have zero-voltage switching capability with minimal circuit components.

BACKGROUND OF THE INVENTION

Transition resonance for bridge-type converters has been described in U.S. Pat. Nos. 4,864,479 and 4,912,622 of R. L. Steigerwald and K.D.T. Ngo, assigned to the present assignee and incorporated by reference herein. Such circuits typically use four power switches (FET's) for high power applications, e.g., 200 Watts and greater. Although there is only one power processing magnetic component, i.e., transformer or inductor, it is impractical to use these circuits with four switches and associated gate drivers for lower power (less than 50 Watts) applications due to the relatively high costs associated therewith. Quasi-resonant and multi-resonant circuits are usually used for lower power conversion at high frequencies with a single power FET. However, although such resonant circuits have only one power FET, they typically have several resonant components (resonant inductors and capacitors) that cause inefficiencies and higher voltage or current stresses on the FET.

Accordingly, it is desirable to provide a dc-to-dc power converter that is amenable to high-density packaging techniques and very high frequency operation (e.g., 1-10MHz). It is further desirable to require only a single main power switch, rather than multiple power switching devices. Still further, it is desirable to achieve zero-voltage switching (i.e., switching with zero-voltage across a switching device) without the need for additional resonant circuit components.

SUMMARY OF THE INVENTION

A dc-to-dc power converter, comprising a transformer, a single primary-side power switching device and a parallel combination of a diode and a capacitance, is controlled to operate in a "natural" zero-voltage switching mode such that the power switching device is switched with zero-voltage thereacross. The converter is controlled such that the voltage across the primary-side power switching device resonates down to zero as the current through the diode resonates to zero before the primary-side power switching device is turned on. In an alternative embodiment, a dc-to-dc power converter is configured for operation in a zero-voltage switching mode at light loads and in a "natural" zero-voltage switching mode at heavier loads. Due to the simplicity of the circuit, i.e., its very few components, and its high-frequency capability, a converter according to the present invention can be implemented using high-density packaging techniques. Such a converter is useful, for example, in ultra-high -density point-of-load dc-to-d.c converters for distributed power systems.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and advantages of the present invention will become apparent from the following detailed description of the invention when read with the accompanying drawings in which:

FIG. 1 schematically illustrates a high-frequency resonant flyback switching cell;

FIG. 2 graphically illustrates ideal current and voltage waveforms for the resonant flyback cell of FIG. 1;

FIG. 3 schematically illustrates a high-frequency resonant flyback switching cell configured for natural zero-voltage switching according to the present invention;

FIG. 4 graphically illustrates ideal current and voltage waveforms for the resonant flyback switching cell of FIG. 3 using natural zero-voltage switching in accordance with the present invention;

FIG. 5 graphically illustrates simulation results for a resonant flyback switching cell, such as that of FIG. 3, operating in the natural zero-voltage switching mode in accordance with the present invention;

FIG. 6 schematically illustrates a high-frequency resonant buck switching cell configured for natural zero-voltage switching according to the present invention;

FIG. 7 schematically illustrates a high-frequency resonant buck switching cell with a tapped-transformer configured for natural zero-voltage switching according to the present invention;

FIG. 8 schematically illustrates a high-frequency resonant Cuk switching cell configured for natural zero-voltage switching according to the present invention; and

FIG. 9 schematically illustrates a high-frequency resonant boost switching cell configured for natural zero-voltage switching according to the present invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates a flyback switching cell 10 comprising a transformer 12 with a primary winding inductance L_(pri) and a secondary winding inductance L_(sec). A primary-side power device (e.g., FET) Q₁ is connected in series with the primary winding L_(pri) across an input dc voltage Ein and an input filter capacitor C_(in). A secondary-side power switching device Q₂ is connected in series with the secondary winding L_(sec). The integral parasitic diodes for switching devices Q₁ and Q₂, respectively, are illustrated in FIG. 1 as D_(Q1) and D_(Q2), respectively. The primary-side current is designated as i_(pri), and the secondary-side current is designated as i_(sec) in FIG. 1. The output dc voltage E_(out) of the switching cell is taken across an output filter capacitor C_(out). FIG. 2 graphically illustrates ideal current and voltage waveforms for operating the converter of FIG. 1 in a zero-voltage switching mode, i.e., switching the power devices Q₁ and Q₂ with zero voltage thereacross. In operation, device Q₁ turns on to build up current in the transformer primary magnetizing inductance (primary winding) L_(pri). Q₁ is then switched off, interrupting the primary inductance current i_(pri). The core energy is then transferred to the secondary as current increases in the secondary inductance (secondary winding) L_(sec). Q₂ is turned on when Q₁ turns off so that the secondary current i_(sec) flows in Q₂ 's channel in the reverse direction. With proper selection of the secondary-side switching device Q₂, i.e., to provide synchronous rectification, the voltage drop across it (in the reverse direction) can be significantly less than that which would occur in a Schottky rectifier, resulting in significantly higher converter efficiency for low output voltage converters, e.g., 5 volts and less. (A synchronous rectifier is a very low on-resistance power FET.) The turns ratio of the transformer 12 is selected so that each FET Q₁ and Q₂ conducts for approximately a 50% duty cycle at the nominal input and output voltages. The output voltage E_(out) is regulated by varying the duty cycle of switches Q₁ and Q₂ in conventional manner.

In the zero-voltage switching mode, energy that is stored in a FET output capacitance (measured from drain to source) when the device-is off is not dissipated within the device when it is turned on, but is resonated back to the dc supply as energy stored in the primary and secondary magnetizing inductance resonates with the output capacitance of the FET to ring the voltage thereacross to zero before the gate is turned on. For example, when Q₁ turns off, the "undotted" terminals 14 and 16 of the transformer windings become positive. This has the effect of reducing Q₂ 's drain voltage V_(Q2) to zero before Q₂ turns on. After the secondary current i_(sec) through Q₂ is driven to zero as the inductance L_(sec) releases its energy, device Q₂ is kept on so that the current momentarily reverses in Q₂, i.e., flows from drain to source from the "dotted" terminal 17 of winding inductance L_(sec). Then Q₂ is turned off, and the energy stored in the core of inductance L_(sec) causes the "dotted" terminals 15 and 17 of the transformer 12 to become positive, driving Q_(l) 's drain voltage V_(Q1) to zero before Q₁ turns on. Thus, Q₁ is turned on with zero voltage thereacross. In this manner, turning off one FET causes the other FET to be switched with zero voltage thereacross. And, by varying the on-time of Q₁, the output voltage can be regulated using a well-known constant frequency PWM control.

Disadvantageously, in the circuit of FIG. 1, there is a trade-off to achieving zero-voltage switching. In particular, some energy is circulated back and forth between the primary and secondary windings of the transformer 12 in order to ring the voltages V_(Q1) and V_(Q2) across the FET's Q₁ and Q₂, respectively, to zero. This necessarily results in somewhat more current in the switches Q₁ and Q₂ than would otherwise be present in a conventional flyback circuit.

In accordance with the present invention, a simple circuit and method of operation are provided which achieves zero-voltage switching with reduced circulating current in the power switches, i.e., which does not require an active reversal of current in the secondary winding. Furthermore, only one power switching device and a diode are required, without additional resonant circuit elements.

FIG. 3 illustrates a high-frequency resonant flyback switching cell 20 suitable for controlling by "natural" zero-voltage switching in accordance with the present invention. In the circuit of FIG. 3, the parallel combination of a diode D₂ and a capacitance C₂ (which may comprise the parasitic capacitance of the diode D₂) replaces the secondary-side switching device Q₂ of FIG. 1.

FIG. 4 graphically illustrates operation of the circuit of FIG. 3 in a "natural" zero-voltage switching mode according to the present invention. As energy is being released from the secondary winding L_(sec), the current i_(sec) in L_(sec) (and D₂) falls toward zero. When the secondary current i_(sec) reaches zero, diode D₂ is immediately off, and inductance L_(sec) resonates with capacitance C₂ (being driven by the output voltage E_(out)), causing the voltage V_(D2) to ring up toward twice the output voltage E_(out). This voltage, transformed by the transformer, is just enough to ring the voltage V_(Q1) on Q₁ 's drain to zero. Hence, Q₁ is zero-voltage switched for the case where the converter is driven at or nearly at a 50% duty cycle.

As shown in FIG. 4, the secondary current i_(sec) is approximately a half-sinusoidal pulse. The secondary voltage V_(D2) rings up (approximately cosinusoidally), and the primary drain voltage rings down (approximately cosinusoidally), before Q₁ is switched on.

When operating in the "natural" zero-voltage switching mode according to the present invention, frequency may be varied to control output voltage, as described hereinabove, because the conduction time of the secondary is not constant, rather it is a function of the load current, i.e., the time for the secondary current i_(sec) to discharge to zero.

To optimize operation at high frequencies, the transformer windings L_(pri) and L_(sec) should be tightly coupled. For example, a suitable transformer may be implemented in a high density interconnect (HDI) structure, such as, for example, of a type described in commonly assigned U.S. Pat. application Ser. No. 08/043,166 of W. A. Roshen, C. S. Korman and W. Daum, filed Apr. 1, 1993 and incorporated by reference herein. HDI technology is described generally in commonly assigned U.S. Pat. No. 4,783,695 of Eichelberger et al, also incorporated by reference herein.

FIG. 5 graphically illustrates simulation results for a resonant flyback switching cell, such as that of FIG. 1, constructed and operated in a "natural" zero-voltage switching mode according to the present invention for a 10MHz, 50 to 10 Vdc, 50 Watt converter.

The circuit of FIG. 1 with primary and secondary devices Q₁ and Q₂ can also be operated in the "natural" zero-voltage switching mode of the present invention by turning off device Q₁ when its drain current reaches zero, rather than leaving it on to store energy as shown in FIG. 2. As a result, the output capacitance of Q₂ will ring up toward twice the output voltage just as the output voltage of diode D₂ as shown in FIG. 4, allowing device Q₁ to operate in a zero-voltage switching mode. Thus, the circuit of FIG. 1 can operate in either the normal zero-voltage switching mode, as described hereinabove with reference to FIG. 2, or the "natural" zero-voltage switching mode according to the present invention, depending upon gating control of Q₂.

When operating in the natural zero-voltage switching mode, the control will increase frequency as the load lightens. At very light loads, frequency may be too high to obtain the desired performance. To avoid problems at very light loads, the switching device Q₂ with its parasitic diode D_(Q2) (FIG. 1) may be used for operation in "dual" modes. That is, at or about a threshold frequency, the circuit of FIG. 1 may be operated in a zero-voltage switching mode, as described hereinabove with reference to FIGS. 1-2, and in a natural zero-voltage switching mode at frequencies below a threshold frequency, as described hereinabove with reference to FIGS. 3-5. To this end, at light loads (i.e., above the threshold frequency), Q₂ is left on to allow a sufficient reversal in current in L_(sec) such that energy builds up in L_(sec) to allow the voltage across Q₁ to resonate to zero, thus allowing for zero-voltage switching of Q₁. On the other hand, at heavier loads (i.e., below the threshold frequency), Q₂ remains off and operation in a natural zero-voltage switching mode-proceeds via the parasitic diode D_(Q2) in the manner described hereinabove. Advantageously, therefore, with this "dual" mode operation, Q₂ can be a lower-rated FET than Q₁ because Q₂ is only needed at light loads.

Other known types of switching cells can be modified to substitute a parallel combination of a diode D₂ and a capacitance C₂ (which may comprise the parasitic capacitance of the diode D₂) and operated in a "natural" zero-voltage switching mode according to the present invention. For example, FIG. 6 illustrates a high-frequency resonant buck switching cell 30 configured according to the present invention to operate in a "natural" zero-voltage switching mode according to the present invention, with a magnetic component comprising an inductance L₁. (Current through L₁ is designated in FIG. 6 as i_(L)). The voltage at point A can ring up to a maximum of twice the output voltage E_(out). Hence, such converters are operated to have an approximately 50% or greater duty cycle (i.e., the ratio of E_(out) to E_(in)), which can be obtained by suitable selection of transformer turns ratios and/or taps. For converters having a greater than 50% step-down in voltage from E_(in) to E_(out), FIG. 7 illustrates a high-frequency resonant buck switching cell arrangement 40 with a tapped inductor L₁ /L₂ (with a primary-side inductance L₁ and a secondary-side inductance L₂ determined by the tap). In the natural zero-voltage switching mode, when the current in the diode D₂ reaches zero, L₂ resonates with the capacitance C₂ to ring the voltage at point A toward twice the output voltage E_(out). The inductor tap ensures that the voltage at point A is transformer-coupled so that the source of Q₁ is driven at least to the input voltage E_(in), with the result that the voltage V_(Q1) across Q₁ reduces to zero and zero-voltage switching occurs.

FIG. 8 illustrates a Cuk switching cell arrangement 50 with the secondary-side switching device replaced with the parallel combination of diode D₂ and capacitance C₂ and operated in a "natural" zero-voltage switching mode according to the present invention. In the converter of FIG. 8, energy is stored in the transformer winding inductance and, to zero-voltage switch Q₁, the switch capacitance rings with L_(sec) which is driven by the dc voltage on C_(out).

FIG. 9 illustrates a boost switching cell arrangement 60 with a tapped inductor L₁ /L₂ and the secondary-side switching device replaced with the parallel combination of diode D₂ and capacitance C₂ and operated in a "natural" zero-voltage switching mode according to the present invention. In this case, when the current in diode D₂ goes to zero, the capacitance C₂ rings with the inductor L₁ /L₂ being driven by the difference between the output and input voltages (E_(out) E_(in)). By proper selection of the inductor tap, the voltage at point A (i.e., at the drain of Q₁) will ring to zero, allowing for zero-voltage switching.

Advantageously, the switching cells constructed and operated in a "natural" zero-voltage switching mode according to the present invention at high frequencies approaching, for example, 10MHz are highly efficient and may be implemented using very high density packaging techniques.

While the preferred embodiments of the present invention have been shown and described herein, it will be obvious that such embodiments are provided by way of example only. Numerous variations, changes and substitutions will occur to those of skill in the art without departing from the invention herein. Accordingly, it is intended that the invention be limited only by the spirit and scope of the appended claims. 

What is claimed is:
 1. A dc-to-dc power converter, comprising:a transformer comprising a primary winding and a secondary winding, said transformer dividing the converter into a higher dc voltage primary side and a lower dc voltage secondary side; a single primary-side power switching device; a parallel combination of a diode and a capacitance situated on said secondary side; and control means for operating said power converter in a natural zero-voltage switching mode without actively reversing current in said secondary winding, thereby minimizing circulating currents in said switching device, such that the voltage across said power switching device resonates down to substantially zero as the current through said diode resonates substantially to zero whereupon said control means turns on said primary-side switching device with substantially zero-voltage thereacross.
 2. The converter of claim 1 wherein said control means switches said primary-side switching device with an approximately 50% or greater duty cycle.
 3. The converter of claim 1 wherein said transformer comprises a tapped inductor.
 4. The converter of claim 1 wherein said transformer comprises an HDI structure.
 5. The converter of claim 1, further comprising frequency control means for regulating the output voltage.
 6. The converter of claim 1 wherein said parallel combination of said diode and said capacitance is implemented using another power switching device.
 7. A dc-to-dc power converter,comprising:a transformer comprising a primary winding said and a secondary winding, said transformer dividing the converter into a higher dc voltage primary side and a lower dc voltage secondary side; a primary-side power switching device; a secondary-side power switching device having an integral parasitic diode; and control means for operating said power converter at frequencies below a threshold frequency in a natural zero-voltage switching mode without actively reversing current in said secondary winding, thereby minimizing circulating currents in said switching device, such that the voltage across said primary-side power switching device resonates down to substantially zero as the current through said parasitic diode resonates substantially to zero whereupon said control means switches said primary-side power switching device with substantially zero-voltage thereacross, said secondary-side power switching device remaining off at frequencies below said threshold frequency, said control means operating said power converter in a zero-voltage switching mode at frequencies above said threshold frequency such that the current is reversed in said secondary-side power switching device until there is sufficient energy storage such that said secondary-side power switching device is turned off to allow the voltage across said primary-side power switching device to resonate down to substantially zero whereupon said control means turn on said primary-side power switching device with substantially zero-voltage thereacross.
 8. The converter of claim 7 wherein said control means switches said primary-side power switching device with an approximately 50% or greater duty cycle.
 9. The converter of claim 7 wherein said transformer comprises a tapped inductor.
 10. The converter of claim 7 wherein said transformer comprises an HDI structure.
 11. The converter of claim 7, further comprising frequency control means for regulating the output voltage. 